PAM4 Analysis and Measurement Considerations Webinar

WEBINAR
PAM4 Analysis and Measurement Considerations
16/14/2017
Mike Hertz
mike.hertz@teledyne.com

About Us: Teledyne LeCroy
o LeCroy founded in 1964 by Walter LeCroy
o Origins are high speed digitizers for particle physics
research
o Teledyne LeCroy corporate headquarters is located in
Chestnut Ridge, NY
o Teledyne LeCroy has the most advanced technology
and widest line of Real-Time digital oscilloscopes
(from 40 MHz to 100 GHz)
o Long History of Innovation in Digital Oscilloscopes
o Teledyne LeCroy became the world leader in protocol
analysis with the purchase of CATC and Catalyst, and
creating a protocol analyzer division based in Santa
Clara, CA.
o In August 2012, LeCroy was acquired by Teledyne
Technologies and was renamed Teledyne LeCroy
o In April 2016 we acquired Frontline Test Equipment
and Quantum Data to add wireless and video to our
protocol analyzer portfolio

About the Presenter
1. Field Applications Engineer with
Teledyne LeCroy in Michigan for over
16 years
2. BSEE from Iowa State University and
an MSEE from the University of Arizona
3. Awarded six U.S. patents for
oscilloscope measurement design
Mike Hertz
Senior Field Applications Engineer
Teledyne LeCroy
mike.hertz@teledynelecroy.com

PAM4 Analysis and Measurement Considerations
Using Oscilloscopes

Agenda
• Quick review: eye patterns, bit rate, baud rate, PAM4
nomenclatures
• PAM4 test configurations
• PAM4 compliance measurements
• PAM4 debug techniques
• Real time and sampling scopes for PAM4
• Questions

PAM4: Pulse Amplitude Modulation 4-Level

Bit rate and baud rate
Bit rate (
𝑏𝑏𝑏𝑏𝑏𝑏𝑏𝑏
𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠
) = Baud rate (
𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠
) ∗ (
𝑏𝑏𝑏𝑏𝑏𝑏𝑏𝑏
)
Bit rate is
#𝑏𝑏𝑏𝑏𝑏𝑏𝑏𝑏
Baud rate is
#𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠
PAM4 encodes 2 bits into each symbol, therefore a 56 Gb/s throughput requires a PAM4 signal at 28 Gbaud.
56 Gb/s is often referred to as “50G” because coding overhead results in a reduced data payload capacity.
PAM4-based Ethernet standards for 100G, 200G and 400G use 2, 4, or 8 lanes of 28 Gbaud PAM4.

Ethernet interfaces and nomenclature
Electrical Optical
Source: Ethernet Alliance
Note that NRZ electrical signaling is used at 10 and 25 Gb/s, and PAM4 is exclusively used for rates 50 Gb/s and higher

Ethernet technologies nomenclature (followed by most of the high-speed Ethernet standards)
400GBASE-CDAUI-8
100GBASE-KR2
Aggregate bit rate of
entire link after coding
overhead is removed Interface type
Number of Lanes
Attachment Unit Interface
Backplane interface
Roman numerals for aggregate bitrate
in Gb/s are appended when followed by
“AUI”
400 Gb/s across 8 lanes,
or 50 Gb/s on each lane
100 Gb/s across 2 lanes,
or 50 Gb/s on each lane
Emerging standards such as 50GBASE-KR and 100GBASE-CR2 achieve 50 Gb/s per lane by using PAM4 signaling.
Ethernet standards which have a speed of 50 Gb/s per lane are using 28 Gbaud PAM4 signaling (includes overhead coding),
with a bitrate of 56 Gb/s (data throughput of 50 Gb/s)

OIF and IEEE
 OIF-CEI56G-*-PAM4, where “*” is a variant:
 OIF-CEI56G-XSR-PAM4, for 0-50mm traces
 OIF-CEI56G-VSR-PAM4, for 125mm host + 25mm
module traces
 OIF-CEI56G-MR-PAM4, for 500mm trace + 1 connector
 OIF-CEI56G-LR-PAM4, for 1000mm trace + 2 connectors
 Two IEEE 802.3 (Ethernet) amendments:
 802.3bs, for 400 GbE (8 lanes of 25G PAM4)
 802.3cd, for 50/100/200 GbE (1/2/4 lanes of 25G PAM4)
IEEE 802.3 bs
IEEE 802.3 cd
 The VSR variant involves most of the test requirements for PAM4 because:
 VSR has normative (compliance) tests which are somewhat unique to it and to XSR
 But it also has informative tests which are measurements that also appear in –MR, -LR
and the IEEE specs
 Most measurements in this presentation refer to –VSR unless specified otherwise
 Most of the measurement concepts can be easily generalized

Test point TP1a
“The output of the Host Compliance Board
(HCB) provides access to the host-to-module
electrical signal (host electrical output) defined at
TP1a.”
Signals from the host Tx are measured after
passing through the host’s PCB trace and a
defined Host Compliance Board (HCB). Signals
at TP1a are expected to be measured using a
standard receiver CTLE for CEI-56G.
The HCB can either be a physical compliance
board, or emulated using S-parameters. A .s4p
file is used for the HCB, and CDR and CTLE are
emulated in the scope.
Device
under
test
HCB

Test point TP4
“The output of the Module Compliance Board
(MCB) provides access to the module to host
electrical signal (module electrical output)
defined at TP4.”
The signal from a module Tx is measured after
passing through the host’s PCB trace and a
defined Module Compliance Board (MCB).
Signals at TP4 are expected to be measured
using a standard receiver CTLE for CEI-56G,
and the MCB can either be a physical
compliance board, or emulated using S-
parameters. A .s4p S-parameter file is used for
the MCB, and, CDR and CTLE are
implemented within the scope software.
Device
under
test
MCB

Test setup for TP0a
“TP0a is defined to be separated from TP0, the ball
of the package performing the host-to-module
transmit function, by 1 dB of PCB attenuation at 14
GHz.”
TP0a does not have a well-defined test setup
because it is not at or near a point of connection in
an actual interface. The goal for TP0a is to connect
as close to the output pin of the Tx chip as possible.
This point is typically accessible for silicon/IP
developers who are working on a development
board.
The exact connection setup and embedding or de-
embedding requirements will vary depending on
what development or evaluation board connections
are available. Signals at TP0a are expected to be
measured without any receiver equalizer emulation,
and VSR tests at TP0a are informative only.

PAM4 Differential Voltage, Common Mode Voltage, Common Mode Noise
 Differential voltage pk-pk = max(Dp - Dn) – min(Dp - Dn)
 Max Common mode voltage = max(Dp + Dn)
 Min Common mode voltage = min(Dp + Dn)
 RMS Common mode noise = sdev(Dp + Dn)

Transition Time Measurement
The Transition Time measurement is described in
the standard as:
“Transition times (rise and fall
times) are defined as the time
between the 20% and 80% times, or 80%
and 20% times, respectively, of
isolated -1 to +1 or +1 to -1 PAM4
edges. Using the QPRBS13-CEI test
pattern the transitions within
sequences of three -1s followed by
three +1s, and three +1s followed by
three -1s, respectively, are
measured. These are PAM4 symbols 1820
to 1825 and 2086 to 2091,
respectively, where symbols 1 to 7
are the run of seven +1’s. In this
case, the 0% level and 100% level may
be estimated as the average signal
within windows from -1.5 UI to -1 UI
and from 1.5 UI to 2 UI relative to
the edge.”
Histogram of rise times
Histogram of fall times Histogram of slew rates

CEI-56G-VSR-PAM4 and 106
 Signaling speed
 Baud rate of 18 – 29 Gbaud
 Most devices operate at 28 Gbaud
 This corresponds to a raw bit rate
of 56 Gb/s
 Forward Error Correction (FEC)
 CEI-56G-VSR-PAM4 only requires
a raw BER of only 10-6 (very
different from NRZ signals)

Eye Diagram measurements and 106 UI
Most of the eye diagram measurements are
made with specific reference to 10-6
contours because the threshold for FEC in
use by these standards is a BER of 10-6 at
the physical layer. Therefore, unlike NRZ
analysis, extrapolation is not needed for
PAM4 analysis, because 106 symbols can
be easily captured in a single acquisition on
a real-time oscilloscope.
Acquire a sufficient number of UI’s, at least
106, and ideally 107.
10-6 contours are displayed here in green.

BER Contour Description
Click here

Eye Diagram measurements: Tmid
Tmid is the midpoint of
the maximum horizontal
eye opening of the 10-3
(red) inner eye contour
of the middle eye.
Tmid is the expected
time position for the
hardware receiver to
sample the signal.
The Tmid calculation is
the starting point for
many other eye-diagram
based measurements.
Tmid

Eye Diagram Measurements: EH6 (Eye Height @ 10-6)
EH6 represents the height of the respective
eye at a BER of 10-6, (green contour)
determined from voltage CDFs in a +/-
0.025 UI time window centered on Tmid.
Tmid
EH6 upp
EH6 mid
EH6 low
Note that EH6 is not necessarily measured at the point of
maximal eye height, since EH6 must be measured at
Tmid, the midpoint of the 10-3 (red contour). EH6 is the
distance between intersection points of Tmid and the 10-6
contour ring for each eye.

Eye Diagram measurements: EW6
EW6 is the width of the respective eyes
determined from CDFs of eye edges
halfway between the 10-6 points of the
voltage CDFs of the middle eye (EH6/2).
Note that EW6 is not necessarily
measured at the point of maximal eye
width (especially observable for the
upper and lower eyes), since the EW6
measurement location is (EH6/2) for
each eye.
(EH6 mid)/2
EW6 upp
EW6 low
EW6 mid
(EH6 upp)/2
(EH6 low)/2

Eye Diagram Measurements: Eye Linearity
Eye Linearity is an alternative to the RLM
measurement (discussed later). Eye
Linearity is defined as:
Where AVupp, AVmid and AVlow are the
average of the eye amplitudes (not heights),
defined as the difference of the mean levels
of the upper and lower level voltage
histograms in a +/- 0.025 UI time window
centered at Tmid.
The measurement of Eye Linearity
determines symmetry of the three eyes. An
ideal signal has an eye linearity of 1.000.
AVupp
AVmid
AVlow

Vertical Symmetry and Eye Linearity
Examples of good eye linearity Examples of problematic eye linearity

Eye Diagram measurements: Mask test
The VSR mask is purely horizontal
and is defined as:
“…an Eye Width mask centered on
Tmid having a width from the relevant
table which extends above and below
the waveform for the upper and lower
PAM4 eyes. The EW6 low, middle and
upper eye edges shall be outside this
Eye Width mask.”
Because the mask is centered on
Tmid, it’s possible for the upper and
lower eyes to pass the EW6 test (wide
enough at 10-6) but fail the mask test
(not sufficiently centered on Tmid)
Mask

Horizontal Symmetry and Mask Testing
Horizontally-symmetrical eyes passing mask test: Asymmetrical eyes failing mask test:

Skew Between Generated Bits Affecting Horizontal Symmetry
Asymmetrical eyes failing mask test:

Background on Jitter Methodology for PAM4
Many NRZ signal standards require extrapolation of Total Jitter (Tj) to BERs
of 10-12. This required fitting values to a model (dual-dirac), and the terms
involved (Tj, Rj, Dj) are associated with those models and extrapolation
methods.
PAM4 technologies require only a BER of 10-6 at the physical layer. Since
oscilloscopes can easily acquire 106 bits in a single acquisition, traditional
methods of Rj/Dj extrapolation are not needed, and new methods and
terminology is used for PAM4 signaling.

Jitter Methodology for PAM4
 UUGJ (Unbounded, Uncorrelated Gaussian Jitter) – Conceptually similar to random jitter
 UBHPJ (Uncorrelated, Bounded High-Probability Jitter) – Conceptually similar to deterministic
jitter
 EOJ (Even-Odd Jitter) – Systematic jitter occurring between even- and odd-numbered
transitions. This was usually called “F/2 jitter” in an NRZ context, and was often mistaken for
DCD.
 UJ4 – Measured peak-peak uncorrelated jitter at the 10-4 probability level
 UJ6 – Measured peak-peak uncorrelated jitter at the 10-6 probability level
 J4 – Measured peak-peak jitter at the 10-4 probability level
 J6 – Measured peak-peak jitter at the 10-6 probability level
Note: J4 and J6 are deprecated terms that were used in jitter calculations before UJ4 and UJ6
were adopted in more recent revisions

Uncorrelated jitter (UJ4 and UJ6)
 A repeating
pattern must
be used
 For each
transition in
the pattern,
form a
histogram of
its edge
arrival times

Uncorrelated jitter (UJ4 and UJ6)
 Remove the
mean from
each edge
histogram
and sum it
with all other
edges from
the same eye
 Now we have
an
uncorrelated
jitter (UJ)
histogram for
each of the
three eyes

Calculating UUGJ and UBHPJ
 A jitter CDF is derived from each
histogram
 J4 and J6 are calculated as the
width of the CDF at 10-4 and 10-6
respectively
 UUGJ and UBHPJ are
calculated from:
*Equation from CEI-56G-MR-PAM4 spec

Even-odd jitter
 “Even-odd jitter is measured using two repetitions of a QPRBS13-CEI
test pattern with FIR off. The deviation of the time of each transition from
an ideal clock at the signaling rate is measured.
Even-odd jitter is defined as the magnitude of the difference between
the average deviation of all even-numbered transitions and the average
deviation of all odd-numbered transitions, where determining if a
transition is even or odd is based on possible transitions but only actual
transitions are measured and averaged.”
(from CEI-56G-MR-PAM4 spec)
 Note QPRBS13-CEI is a pattern with an odd number of symbols – so in
any repetition, each symbol will land on the “opposite” even/odd clock
edge than it did in the previous repetition.

Linear Fit Method
1. Acquire waveform on scope.
2. Resample the waveform using an integer number of samples.
3. Generate an ideal waveform with the same pattern.
4. Deconvolve to obtain a pulse response of a full-swing 0-to-3-to-0 transition to
determine the impulse response of the system.
The mathematical definition is described in
OIF-CEI-03.1, section 11.3.1.6.4.
The linear fit error is calculated as the
difference between the pulse response and the
actual signal for each resampled point.

Linear fit pulse response - examples
 This signal is
very clean
 Note how
controlled the
pulse
response is
 The impulse
response is
derived from
the waveform.
Linear fit pulse response

 This signal
has twice as
much noise.
 Note the pulse
response
hasn’t
changed.

 This signal has
low noise but is
severely
bandwidth-
limited.
 Note the pulse
response
reflects the
signal shape.

 Since LFPR is sampled an integer
number of times per UI, we can
decimate it to get one amplitude
value per UI.
 This can be used to optimize
transmitter emphasis coefficient
values.
 Removes all noise, all pattern
dependent artifacts to produce
ideal normalized coefficient values.
decimated to one point per UI

SNDR
The linear fit error is the difference between the
actual transmitter output signal and the ideal signal,
producing an error vector e(k). SNDR is calculated
using the maximum value of the pulse peak, pmax,
and the linear fit error, e(k). Note the RMS
deviations of the voltage levels are not used in the
calculation.
*from CEI-56G-MR-PAM4 spec
Signal-to-noise-and-distortion ratio (SNDR)
is calculated using the linear fit pulse
response and the linear fit error:

Transmitter Linearity (RLM)
 This measurement is not required in
normative or informative –VSR tests, but
is required by most other variants
 It is conceptually similar to Eye Linearity
but is measured in a different way (and
the resulting values are not directly
comparable)
 A perfect signal has an RLM of 1 but it
does not scale the same as eye linearity
 The definitions of the signal levels V-1, V-
1/3, V+1/3, V+1 have changed as the
standard evolved
 RLM is also referred to as “Level
Separation Mismatch Ratio”

RLM – the “old” way
 In older specifcation revisions, RLM
required a special pattern to be
used.
 Note in this pattern, each symbol is
repeated for 16 UIs to ensure
settling.
 The level values were determined
as the voltage at the middle of the
run of 16 symbols.
 This pattern was difficult for many
device vendors to generate.
*from IEEE p802.3bj

RLM – the “new” way
 Now as described in the –VSR spec (and referenced by the others), RLM
derives the voltage levels directly from a QPRBS13 signal (which is
much easier to generate)
 The voltage level at the center of each symbol in the pattern is
measured, and these are used to calculate a mean value for each level.

RLM – what is measured currently
 The RLM calculation was developed
when the “old” way was in use, but
was generalized to estimate RLM for
arbitrary patterns:
 Find the longest run of each level
(must be >6 symbols)
 Use the center point of this run to
calculate the voltage value for that
level
 The result is the correct calculation
on the (now-deprecated) special
pattern, and good on other patterns
 Considerations:
 If the pattern does not have a
run of >6 UI of all 4 levels, no
values are produced
 The measurement can change
substantially if the pattern
changes
 Long patterns with long
consecutive symbols yield the
best results

RLM variation as a function of pattern length
PRBS7 PRBS13

PAM4 multiple eye pattern view

Fixture de-embedding with S-parameters, virtual probing, and receiver equalization
Physically
probed
Virtually
probed
Virtually
probed
Equalized

Intersymbol Interference Measurement on PAM4

Data-Dependent Jitter Plot on PAM4

Oscilloscope Bandwidth Selection:
Power Spectral Density Example of 28 Gb/s SERDES
The Power Spectral Density of a 28 Gb/s serial data waveform is plotted above, with Power (dB) on the Y-axis and Frequency (GHz) on the X-
axis. For a 28 Gb/s signal: the fundamental frequency is centered at 14 GHz, there is a null at 28 GHz, the third harmonic is centered at 42
GHz, and the next null is at 56 GHz. Therefore, an oscilloscope with at least 56 GHz bandwidth is needed in order to capture all of the power
spectral density of the third harmonic of a 28 Gb/s signal, and all of the PSD associated with the third harmonic will be captured by a
LabMaster 10-60Zi 60 GHz oscilloscope. The darker blue area is the extra power spectral density provided by a 60 GHz oscilloscope
compared to a 32 GHz oscilloscope.
Illustration of harmonic content forming a bit pattern

Impact of Bandwidth Reduction on a PAM4 Signal

Recommended Oscilloscope Hardware and Software for PAM4 Testing
Serial Data Analyzer oscilloscope
(probably 65 GHz bandwidth)
PAM4 compliance test software
for conformance (and debug)
PAM4 signal analysis software
for PAM4 debug

Real Time and Sampling Scopes
Realtime Scope
A realtime scope (bandwidths up to 100 GHz)
typically triggers on a waveform event, then
collects many sample points (millions, billions)
from the single trigger event. The sample
resolution of a realtime acquisition can be as
low as 4.16 ps/pt, which is the inverse of the
sample rate (up to 240 GS/s)
Sampling Scope
A sampling scope (bandwidths up to 80 GHz)
typically triggers on a reference clock, then
collects one sample point per trigger. The
typical sampling scope maximum sample rate is
200 kS/s (slower for long patterns).
• Teledyne LeCroy has the world's highest bandwidth realtime
scope (100 GHz) with world’s highest sample rate (240 GS/s)
Sampling oscilloscope
Bandwidths up to 80 GHz
Max sample rate: 200 kS/s (very
undersampled)
Not able to capture one-time events
Events must be repetitive
Trigger source is mandatory
No software clock recovery for jitter
Limited debug and analysis
Real time oscilloscope
Bandwidths up to 100 GHz real time
Max sample rate: 240 GS/s real time
Able to capture one-time events:
transients, runts, glitches, etc.
Trigger source not required
Software clock recovery for jitter
Advanced debug and analysis

Real Time Scopes and Sampling Scopes for PAM4 Testing
 A real time scope is able to sample at rates up to 240 Gigasamples per second real time, while a
sampling scope is limited to undersampling the signal at a maximum rate of approximately 200
Kilosamples per second (one million times slower than a real time scope). Unlike a real time
scope, a sampling scope cannot capture a contiguous block of data, so the risk is that low-
frequency anomalies cannot and will not be detected or measured.
 A real time scope does not require an external clock source, which completely eliminates trigger
jitter from the measurement. A sampling scope always uses an external clock, which adds jitter
that is not real, into the measurement.
 A real time scope is able to detect periodic jitter on contiguous waveform edges and demodulate it,
then display the modulation profile. Unlike a real time scope, a sampling scope cannot capture
contiguous data and therefore cannot identify the source of jitter.
 A real time scope is able to display the periodic jitter spectrum in the frequency domain. A sampling
scope cannot do this. By displaying the periodic jitter spectrum, the frequency content of jitter
sources is revealed.
 A real time scope can detect one-time glitches in a PAM4 signal. Undetected glitches can cause bit
errors at the receiver. A sampling scope cannot detect one-time glitches.

 A real time scope can detect one-time runts in a PAM4 signal. Undetected runts can cause bit
errors at the receiver. A sampling scope cannot detect one-time runts.
 A real time scope can detect one-time non-monotonic edges in a PAM4 signal. Undetected non-
monotonic edges can cause bit errors at the receiver. A sampling scope cannot detect one-time
non-monotonic edges.
 A real time scope can detect a signal dropout in a PAM4 waveform. Signal dropouts can result in
system malfunction. A sampling scope cannot detect a signal dropout.
 A real time scope is able to isolate intersymbol interference (ISI) and display the effects of ISI on
the PAM4 pattern. A sampling scope does not have an ISI plot, and therefore cannot determine the
effect of bit order on jitter independently of the serial data pattern.
 A real time scope is able to isolate data dependent jitter (DDj) and display the effects of DDj on the
PAM4 pattern. A sampling scope does not have a DDj plot, and therefore cannot determine the
effect of bit order on jitter relative to the serial data pattern.
 A real time scope can generate a jitter simulation signal in software, used to verify PAM4 setups
without the need for a physical signal. A sampling scope cannot do this.

 Not all devices under test have access to the clock signal. In this case, a sampling scope cannot
be used since it requires access to the clock signal. A real time scope does not require a clock
signal.
 A real time scope implements a software clock recovery, completely eliminating the effects of CDR
jitter. A sampling scope does not use a software CDR; it uses a hardware CDR, always introducing
hardware CDR jitter.
 A sampling scope cannot be used for debug in the event of a PAM4 compliance test failure. A real
time scope provides detailed analysis capabilities including advanced math, measurements, and
custom algorithms to identify root cause of failure.
 A real time scope allows for simplified deskewing of a differential signal pair using an automated
process. A sampling scope often requires expensive phase adjusters and a tedious manual
deskewing process.
 A real time scope can perform timing measurements between data and clock while viewing both
waveforms, and can analyze the system clock. Since a sampling scope requires using the same
system clock as the reference, it cannot detect problems with the system clock. This is important
to identify and troubleshoot clock-related problems.
 A real time scope is able to generate an eye pattern up to 1,000 times faster than a sampling
scope allowing for faster accumulation of meaningful, statistical data. Testing PAM4 using a real
time scope results in faster time to insight.

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