Chapter4

CHAPTER4 Linear Wire Antennas
4.1 INTRODUCTION .............................................................................................................................................................................................................................. 2
.
4.2 INFINITESIMAL DIPOLE ................................................................................................................................................................................................................... 2
4.2.1 Radiated Fields ..................................................................................................................................................................................................................... 3
4.2.2 Power Density and Radiation Resistance ............................................................................................................................................................................ 7
4.2.3 Near‐Field ( ) Region .............................................................................................................................................................................................. 13
4.2.5 Intermediate‐Field (kr > 1) Region ..................................................................................................................................................................................... 15
4.2.6 Far‐Field (kr >> 1) Region ................................................................................................................................................................................................... 17
4.2.7 Directivity ........................................................................................................................................................................................................................... 19
4.3 SMALL DIPOLE ....................................................................................................................................................................................................................... 21
4.4 REGION SEPARATION .............................................................................................................................................................................................................. 25
4.4.1 Far‐Field (Fraunhofer) Region ............................................................................................................................................................................................ 27
4.4.2 Radiating Near‐Field (Fresnel) Region ............................................................................................................................................................................... 30
4.4.3 Reactive Near‐Field Region ................................................................................................................................................................................................ 32
4.5 FINITE LENGTH DIPOLE ................................................................................................................................................................................................................. 33
4.5.1 Current Distribution ........................................................................................................................................................................................................... 33
4.5.2 Radiated Fields: Element Factor, Space Factor, and Pattern Multiplication ..................................................................................................................... 35
4.5.3 Power Density, Radiation Intensity, and Radiation Resistance ......................................................................................................................................... 37
4.5.4 Directivity ........................................................................................................................................................................................................................... 41
4.5.5 Input Resistance ................................................................................................................................................................................................................ 42
.
4.6 HALF‐WAVELENGTH DIPOLE ......................................................................................................................................................................................................... 45
4.7 LINEAR ELEMENTS NEAR OR ON INFINITE PERFECT CONDUCTORS ............................................................................................................................................. 49
4.7.1 Image Theory ..................................................................................................................................................................................................................... 50
4.7.2 Vertical Electric Dipole ....................................................................................................................................................................................................... 53
1. Radiation pattern ................................................................................................................................................................................................................ 54
2. Radiation power and directivity ......................................................................................................................................................................................... 57
.
3. monopole ............................................................................................................................................................................................................................ 61
4.7.4 Antennas for Mobile Communication Systems ................................................................................................................................................................. 63
4.7.5 Horizontal Electric Dipole .................................................................................................................................................................................................. 67
PROBLEMS .......................................................................................................................................................................................................................................... 74

4.1
1 INTROD
DUCTION
Wire antennas,
a , linear or curved, are some e of the o
oldest, sim
mplest, cheapest,
an
nd the mo ile for many applica
ost versati ations.
4.2
2 INFINITESIMAL D
DIPOLE
 Infinitesimal dipol
les are no
ot practica
al, they ar
re used to
o represen
nt capacit
tor‐plate
antennas
s.
 In additio
on, they a
are utilized as build
ding more
e complex
x geometr
ries.

The end pl lates are used to
e
provide c e loading to mainta
capacitive ain
the current on the dippole neaarly
uniform.

The plates are very small, their radiation is usually negligible. The wire, in
addition to being very small (l <<), is very thin ( ). The spatial variation of
the current is assumed to be constant
′ ; = constant (4‐1)
4.2.1 Radiated Fields
To find the fields radiated by the current element, it will be required to
determine first and and then find the and .
1. Calculation of
Since the source only carries an electric current , therefore and the
potential function are zero. To find we write

, , ′ ′ (4‐2)

x, y, z : the observation point ; x’, y’, z’ : the source coordinates
: the distance from any point on the source to the observation point
path C : is along the length of the source

Fo
or the problem of F
Figure 4.1

, , 4‐3
0 (infinite
esimal dip
pole)

′

so
o we can w
write (4‐2) as
/
, , /
(4‐4)

2. Calculation of and
To calculate and , it is simpler to transform (4‐4) from rectangular to
spherical components.

(4‐5)
0

For this problem, 0, so (4‐5) using (4‐4) reduces to

(4‐6)
0

⟹ (4‐7)

Substituting (4‐6) into (4‐7) reduces it to
0
(4‐8)
1

Th
he electric
c field E ca
an now be
e found. T
That is,

∙ (4‐9)

1

1 (4‐10)
0

The and ‐field components
are valid ev
e verywher except on the
re, t
so
ource itself, and th are sketched
hey s
in Figure 4
4.1(b) on the surfa of a
ace
sp
phere of ra adius .

4.2.2 Power Density and Radiation Resistance
The input impedance of an antenna consists of real and imaginary parts. For a
lossless antenna, the real part of the input impedance was radiation resistance.
To find the input resistance for a lossless antenna, the following procedure is
taken.

For the infinitesimal dipole, the complex Poynting vector can be written using
(4‐8a)–(4‐8b) and (4‐10a)–(4‐10c) as
1 ∗
1 ∗

2 2
∗ ∗
(4‐11)

1
⟹ (4‐12)
| |
1

Since  is imaginary, it will not contribute to real radiated power. The
reactive power density, which is most dominant for small values of , has both
radial and transverse components. It merely changes between outward and inward
directions to form a standing wave at a rate of twice per cycle. It also moves in the
transverse direction.
The complex power moving in the radial direction is obtained by integrating
(4‐11)–(4‐12b) over a closed sphere of radius r. Thus it can be written as

∯ ∙ ∙ 4‐13

⟹ 1 (4‐14)

Equation (4‐13), which gives the real and imaginary power that is moving
outwardly, can also be written as

∗
∙ 1 P j2ω W W (4‐15)
Where: P power in radial direction ; Prad time‐average power radiated
W time‐average magnetic energy density in radial direction
W time‐average electric energy density in radial direction
2 W W time‐average imaginary reactive power
From (4‐14)

P ; 2ω W W (4‐16, 17)

It is clear from (4‐17) that When kr ∞, the reactive power diminishes
and vanishes.

1. radiation resistance of the infinitesimal dipole
Since the antenna radiates its real power through the radiation resistance, for
the infinitesimal dipole it is found by equating (4‐16) to

| | ⇒ 80 (4‐18, 19)

For a wire antenna to be classified as an infinitesimal dipole, its overall length

must be very small (usually ).

Example 4.1
Find the radiation resistance of an infinitesimal dipole whose overall length is
/50.
Solution:
Using (4‐19)
1
80 80 0.316
50
Since the radiation resistance of an infinitesimal dipole is about 0.3 ohms, it
will present a very large mismatch when connected to practical transmission lines,
many of which have characteristic impedances of 50 or 75 ohms. The reflection
efficiency ( ) and hence the overall efficiency ( ) will be very small.

2. The reactance of an infinitesimal dipole is capacitive.
This can be illustrated by considering the dipole as a flared open‐circuited
transmission line. Since the input impedance of an open‐circuited transmission line
a distance from its open end is given by

2
where is its characteristic impedance, it will always be negative (capacitive) for
≪ .

4.2.3 Near‐Field ( ) Region
An inspection of (4‐8) and (4‐10) reveals that for   / 2 they
can be approximated by

(4‐8a, 10c)  (4‐20c)

(4‐8b)  (4‐20d)

(4‐10a)  (4‐20a)

(4‐10b)  (4‐20b)

 The E‐field components, and are in time‐phase;
 They are in time‐phase quadrature with the H‐field component  ;
 Therefore there is no time‐average power flow associated with them. This is
demonstrated by forming the time‐average power density as
∗ ∗
W Re E H∗ Re

| |
⟹W Re 0 (4‐22)

Equations (4‐20a) and (4‐20b) are similar to those of a static electric dipole and
(4‐20d) to that of a static current element. Thus we usually refer to (4‐20a)–(4‐20d)
as the quasi‐stationary fields.

4.2.5 Intermediate‐Field (kr > 1) Region
As the values of begin to increase and become greater than unity, the
terms that were dominant for ≪ 1 become smaller and eventually vanish.

1 (4‐8b)  (4‐23d)

1 (4‐10a)  (4‐23a)

1 (4‐10b) (4‐23b)

For moderate values of :
 The E‐field components lose their in‐phase condition and approach
time‐phase quadrature.
 Their magnitude is not the same, they form a rotating vector whose
extremity traces an ellipse. This is analogous to the polarization problem
except that the vector rotates in a plane parallel to the direction of
propagation and is usually referred to as the cross field.

 At these intermediate values of , the and components
approach time‐phase, which is an indication of the formation of
time‐average power flow in the outward direction.

(4‐8a, 10c)  (4‐23c)

(4‐8b)  (4‐23d)

(4‐10a)  (4‐23a)

(4‐10b) (4‐23b)

The total electric field is given by
(4‐24)

4.2.6 Far‐Field (kr >> 1) Region
In a region where ≫ 1 , (4‐23a) – (4‐23d) can be simplified and
approximated by

(4‐8a, 10c)  (4‐26b)

(4‐8b)  (4‐26c)

(4‐10a)  (4‐26b)

(4‐10b) (4‐26a)

The ratio of to is equal to

Z (4‐27)

The E‐ and H‐ field components are perpendicular to each other, transverse to
the radial direction of propagation. The fields form a Transverse ElectroMagnetic
(TEM) wave，its wave impedance is the intrinsic impedance of the medium.

Example 4.2
For an infinitesimal dipole determine and interpret the vector effective length.
At what incidence angle does the open‐circuit maximum voltage occurs at the
output terminals of the dipole if the electric‐field intensity of the incident wave is
10 mV/m? The length of the dipole is 10 cm.
Solution:
Using (4‐26a) and the effective length as defined by (2‐92), we can write that

4 26a
⟹

2 92

The maximum value occurs at  90 and it is equal to . The open‐circuit
maximum voltage is equal to
| ∙ | 10 10 ∙ | 10 volts

4.2.7 Directivity
The real power P radiated by the dipole was found in Section 4.2.2, as
given by (4‐16). The same expression can be obtained by first forming the average
power density, using (4‐26a)–(4‐26c). That is,

∗
Re | | (4‐28)

Integrating (4‐28) over a closed sphere of radius r reduces it to (4‐16).
Associated with the average power density of (4‐28) is a radiation intensity U which
is given by

| | ⟹ (4‐29, 30)

Using (4‐16) and (4‐30), the directivity reduces to

4 (4‐31)
and the maximum effective aperture to
(4‐32)

4.3 SMALL DIPOLE
3 E
 The creation of th he current distribution on a
a thin wiree was disscussed in
n Section
1.4, and it was illu
ustrated w
with somee examplees in Figure 1.16.
 The radiaation propperties of
f an infinit
tesimal di
ipole were
e discusseed in the previous
section. I
Its curren
nt distribution was assumed to be con nstant.
A consttant curre
ent distrib
bution is n
not realizable. A be
etter approximatioon of the
cu
urrent disttribution o ntennas, (/50
of wire an ( /10) is the tr
riangular v
variation,
whhich is sho
own in Fig
gure 4.4(bb)

1 , 0
, ,
1 , 0
(4
4‐33)

Th
he vector potential can be w
written using (4‐33) as
/
/
1 1 (4‐34)

Becausse the lenngth of th
he dipole is very sm
mall /10 , for diff
ferent ’
alo wire are not much different from . T
ong the w Thus c
can be ap
pproximatted by
throughoutt the integ
gration pa
ath.

The maximum phase error in (4 4‐34) by allowing will be /2
g
/
/10 18 ffor /10. Thiss amount of phase error hass very litt
tle effect
on rall radiation characteristics. Then, (4
n the over 4‐34) redu
uces to (4‐
‐35)

(4‐35)

which is one‐half of that for the infinitesimal dipole.
/
Ref: , , /
(4‐4)

The potential function (4‐35) becomes a more accurate approximation as kr → ∞.
Since the potential function for the triangular distribution is one‐half of the
corresponding one for the constant (uniform) current distribution, the
corresponding fields of the former are one‐half of the latter. Thus we can write the
E‐ and H‐fields radiated by a small dipole as

(4‐26b)  (4‐36b)

(4‐26a)  (4‐36a)

(4‐26c)  (4‐36c)

Since the directivity of an antenna is controlled by the relative shape of the
field or power pattern, the directivity, and maximum effective area of this antenna
are the same as the ones with the constant current distribution given by (4‐31) and
(4‐32), respectively.
Using the procedure established for the infinitesimal dipole, the radiation
resistance for the small dipole is

80 (4‐18) | |
20 (4‐37)

The small dipole its radiated power is of (4‐18). Thus the radiation
resistance of the antenna is strongly dependent upon the current distribution.

4.4 REGION SEPARATION
Before solving the fields radiated by a finite dipole of any length, it is desirable
to discuss the separation of the space surrounding an antenna into three regions
 The reactive near‐field
 The radiating near‐field
 The far‐field
To solve for the fields efficiently, approximations can be made to simplify the
formulation. The difficulties in obtaining closed form solutions that are valid
everywhere for any practical antenna stem from the inability to perform the
integration of

, , ′ ′ (4‐2, 38)

where
(4‐38a)
In the calculations for infinitesimal dipole and small dipole. The major
simplification of (4‐38) will be in the approximation of R.

The Fig
gure showws a very tthin dipol
le of finite
e length l
l symmetrically pos
sitioned.
Be
ecause the
e wire is v (x’ y’ 0), we
very thin ( e can writte (4‐38) a
as
(4‐39)

wh
hich can b
be written
n as

2

2 ′ (4‐40
0)

Us
sing the binomial e
expansion, we can w
write (4‐4
40) in a se
eries as

⋯ (4‐41)

wh
hose higher order t
terms bec
come less
s significan
nt provide
ed r >> z’.
.

4.4.1 Far‐Field (Fraunhofer) Region
The most convenient simplification of (4‐41) is to approximate it by
≃ ′ (4‐42)

To maintain the maximum phase error of an antenna equal to or less than /8
rad (22.5 ), the observation distance r must equal or be greater than 2 /.
2 / (4‐45)

The usual simplification for the far‐field region is
≃ for phase terms
(4‐46)
≃ for amplitude terms 1/

Ref: , , ′ ′ (4‐38)

For any other antenna whose maximum dimension is , the approximation of
(4‐46) is valid provided
r 2D /λ (4‐47)
For an aperture antenna the maximum dimension is taken to be its diagonal.

It wou
uld seem that thee approxiimation o R in (4‐46) fo
of or the am
mplitude
is more sev
vere than that fo
or the pha
ase.
Ex
xample 44.3
For an antenn with an overall lengt
n na th 5, the o
observations are
made at 60. Find the
e errors in phase and ampplitude ussing (4‐4
46).
So
olution:
For  90 , z’
, 2
2.5, and r
d 6
60, (4‐40
0) reduce
es to

2 2 ′ (4‐40)

 60 2.5 60.052
≃ for phase terms
With (4‐46)
r 60
Therefore the phase difference is
2
∆ ∆ 0.327 18.74 22.5

The difference of the inverse values of R is
1 1 1 1 1 1.44 10

60 60.052
which should always be a very small value in amplitude.

4.4.2 Radiating Near‐Field (Fresnel) Region
If the observation point is chosen to be smaller than 2 / , the maximum
phase error by the approximation of (4‐46) is greater than /8 rad (22.5o).
≃ for phase terms
(4‐46)

If it is necessary to choose observation distances smaller than 2 / , another
term (the third) in the series solution of (4‐41) must be retained to maintain a
maximum phase error of /8 rad (22.5o).

⋯ (4‐41)

Doing this, the infinite series of (4‐41) can be approximated by

(4‐48)

A value of greater than that of (4‐52a) will lead to an error less than /8 rad
(22.5o).

0.385 or 0.62 / (4‐52, 4‐52a)
√ √

The region where the first three terms of (4‐41) are significant, and the
omission of the fourth introduces a maximum phase error of /8 rad (22.5o), is
defined by
/
2 0.62 / (4‐53)

This region is designated as radiating near field because
 The radiating power density is greater than the reactive power density
 The field pattern is a function of the radial distance r.
This region is also called the Fresnel region because the field expressions in
this region reduce to Fresnel integrals.

4.4.3 Reactive Near‐Field Region
If the distance of observation is smaller than the inner boundary of the Fresnel
region, this region is usually designated as reactive near‐field with inner and outer
boundaries defined by
0.62 / > 0 (4‐54)
In summary, the space surrounding an antenna is divided into three regions
whose boundaries are determined by
Reactive near‐field 0.62 / > 0 (4‐55a)
/
Radiating near‐field (fresnel) 2 0.62 / (4‐55b)
/
Far‐field (fraunhofer) 2 0.62 / (4‐55c)

4.5 FINITE LENGTH DIPOLE
The techniques developed previously can be used to analyze the radiation
characteristics of a linear dipole of any length. To reduce the mathematical
complexities, it will be assumed that the dipole has a negligible diameter.

4.5.1 Current Distribution
For a very thin dipole (ideally zero diameter), the current distribution can be
written, to a good approximation, as

, 0
0, 0, (4‐56)
, 0

This distribution assumes that the antenna is
 center‐fed
 the current vanishes at the end points.
 Experiments have verified that the current in a center‐fed wire antenna has
sinusoidal form with nulls at the end points.

For /2 an /2
nd  t the current distrib
bution of (4‐56) is shown
f s
plo
otted in F
Figures 1.1
16(b) and
d (c), respeectively. T
The geommetry of th
he antenn na is that
shown in Figure 4.5.

4.5.2 Radiated Fields: Element Factor, Space Factor, and Pattern
Multiplication
Since closed form solutions, which are valid everywhere, cannot be obtained
for many antennas, the observations will be restricted to the far‐field region.
The finite dipole antenna is subdivided into a number of infinitesimal dipoles
of length ’. For an infinitesimal dipole of length dz’ positioned along the z‐axis
at z’, the electric and magnetic field components in the far field are given as
, ,
(4‐26a)  ′ (4‐57a)

(4‐26b)   (4‐57b)

, ,
(4‐26b)  ′
(4‐57c)
where R is given by (4‐39) or (4‐40).
Using the far‐field approximations given by (4‐46), (4‐57a) can be written as

, ,
′ (4‐58)

Summing the contributions from all the infinitesimal elements to integration. Thus
/ /
/ /
, , ′ (4‐58a)

 The factor outside the brackets is designated as the element factor
 And that within the brackets as the space factor.
For this antenna, the element factor is equal to the field of a unit length
infinitesimal dipole located at a reference point. The total field of the antenna is
equal to the product of the element and space factors.
For the current distribution of (4‐56), (4‐58a) can be written as

′
4 / 2
/
′ ′ (4‐60)

⇒ (4‐62a)

The total component can be written as

(4‐62b)

4.5.3 Power Density, Radiation Intensity, and Radiation Resistance
For the dipole, the average Poynting vector can be written as
∗
∗ ∗

| |
| | (4‐63)

and the radiation intensity as

| |
(4‐64)

The normalized elevation power patterns, for /4, /2, 3/4, and  are
shown in Figure 4.6. The current distribution of each is given by (4‐56). The power
patterns for an infinitesimal dipole ≪    is also included for
comparison.

It is found that the 3‐dB 0
0 330 30
beamwidth of each is equal to
-10
≪ : 3dB beamwidth 90
300 60
-20
/4: 3dB beamwidth 87
/2: 3dB beamwidth 78 -30

3/4: 3dB beamwidth 64 -40 270 90

: 3dB beamwidth 47.8 -30

As the length of the antenna -20
240 120
increases, the beam becomes -10

narrower. Because of that, the 0 210 150
180
directivity should also increase with
1/50 1/4 1/2
length. 3/4 1

As the dipole’s length increases beyond one wavelength  , the number
of lobes begin to increase. The normalized power pattern for a dipole with
1.25 is shown in Figure 4.7.

 Figure 4.7(a) is the
e three‐di
imensiona
al pattern
n
 Figure 4.7(b) is the
e two‐dim
mensional pattern
The cuurrent di istribution for th dipole with
n he es
/4, /2, , 3/ and 2, as given by (4
/2, 4‐56), is
shown in Figure 4.8.
0
0 330 30
-1
10
300 60
-2
20

-3
30

-4 270
40 90

-3
30

-2
20
240 120
-1
10

0 210 150
180 Figure 4.8 Current dist
tributions
Fig
gure 4.7 Thr
ree‐ and twoo‐dimensionnal amplitudde patterns f ength of a li
for a thin along the le inear wire
 and sinuso
dipole of l = 1.25 t distribution.
oidal current antenna.

To find the total power radiated the average Po
d r d, oynting ve
ector of (4‐63) is
int
tegrated o
over a sphhere of ra
adius r. Th
hus

∯ ∙ ∮ ∙

| |
∮ (4‐66)

After some extensive mathematical manipulations, it can be reduced to
| | 1
2 2
4 2
/2 2 2 (4‐68)

where C 0.5772 (Euler’s constant) and Ci x and Si x are the cosine and
sine integrals given by

; 4 68a, b

The radiation resistance can be obtained using (4‐18) and (4‐68)
2 1
2 2
| | 2 2
/2 2 2 (4‐70)

4.5.4 Directivity
The directivity was defined mathematically by (2‐22), or
, |
4 (4‐71)
,

where F ,  is related to the radiation intensity U by (2‐19), or
, (4‐72)
From (4‐64), the dipole antenna of length has

| |
F θ, ϕ F θ , B η (4‐73,73b)

Because the pattern is not a function of , (4‐71) reduces to
|
(4‐74)
,

The corresponding values of the maximum effective aperture are related to
the directivity by

(4‐76)

4.5.5 Inpu
ut Resista
ance
The inp
put imped
dance was defined
d as“the ratio of t
the voltag
ge to curr
rent at a
pa of term
air minals or the ratio of the appropri
r iate comp
ponents of the ele
o ectric to
ma
agnetic fie
elds at a p
point.”
The reaal part of the input
t impedan nce was deefined as the input
t resistanc
ce which
for a lossles
ss antenna reduces adiation resistance.
s to the ra

Th radiati
he ion resist
tance of a dipole of leng l with
e gth
sin current distribution
nusoidal c n is expres
ssed by (4
4‐70).
2

| |
1
2 2
2 2
/2 2 2 (4‐70
0)

By the definition
n, the rad
diation resistance i
is referred
d to the m
maximum
m current
whhich for so /4, 3/4, , etc.) do
ome lengths (l = / he input terminals
oes not occur at th
of the antennna.

To refe
er the radiation res
sistance to
o the inpu ut terminals of
the antenna, the ant tenna is f
first assum
med to be e lossless (RL =
0). Then th power at the in
he nput term
minals is e
equated to the
o
po
ower at th he currennt maximu um. Refer rring to Figure 4.10 0, we
can write
| | | |
⟹ (4‐77)
Figure 4.10 Current
here
wh distribution, m
maximum
does not occcur at the
R rad
diation re
esistance a
at input (f
feed) term
minals
minals.
input term
R = ra
adiation resistance
e at curren
nt maximu
umEq. (4‐
‐70)
I = cu
urrent maximum
I = cu
urrent at input term
minals
dipole of length l, the curre
For a d e input terminals (I ) is re
ent at the elated to

the current maximum
m (I ) ref
ferring to Figure 4.10, by
(4‐78)
he input ra
Thus th adiation r
resistance a) can be written a
e of (4‐77a as
(4‐79)

gure 4.9 R
Fig Radiation resistanc
ce, input r
resistance
e and directivity of a thin dip
pole with
sinu
usoidal cu
urrent distribution.
.

4.6 HALF‐WAVELENGTH DIPOLE
One of the most commonly used antennas is the half‐wavelength (l = /2)
dipole. Because
 Its radiation resistance is 73 ohms very near the 50/75‐ohm characteristic
impedances of some transmission lines,
 Its matching to the line is simplified especially at resonance.
The electric and magnetic field components of a half‐wavelength dipole can be
obtained from (4‐62a) and (4‐62b) by letting l = /2.

, (4‐84, 85)

The time‐average power density and radiation intensity can be written,
respectively, as

| | | |
(4‐86)

| | | |
(4‐86)

Figure 4
4.6 and 4.11 show the two‐ and the t
three‐ dim
mensional
l pattern.
0
0 330 30
-10
0
300 60
-20
0

-30
0

-40
-40 270
0 90

-30
0

-20
0
240 120
-10
0

0 210 150
180

Th
he total po
ower radiated can be obtain
ned as a special cas
se of (4‐67
7)

| |
(4‐88)
| | | |
2 (4‐89)

By
y (4‐69)
2 0.577
72 
ln 2 2
2 0.5
5772 1.838 0.02  2
2.435
(4‐90)

Using (4‐87), (4‐89) and (4‐90), the maximum directivity of the half‐wavelength
dipole reduces to
| /
4 4 1.643 (4‐91)
.

The corresponding maximum effective area is equal to

1.643 0.13 (4‐92)

and the radiation resistance, for a free‐space medium ( 120), is

| |
2 30 2.435 73 (4‐93)
The radiation resistance of (4‐93) is also the radiation resistance at the input
terminals (input resistance) since the current maximum for a dipole of /2
occurs at the input terminals. As it will be shown later, the imaginary part
associated with the input impedance of a dipole is a function of its length (for
/2, it is equal to j42.5). Thus the total input impedance for /2 is equal
to
73 42.5 (4‐93a)

To reduce the imaginary part of the input impedance to zero, the antenna is
matched or reduced in length until the reactance vanishes. The latter is most
commonly used in practice for half‐wavelength dipoles.
 Depending on the radius of the wire, the length of the dipole for first
resonance is about 0.47 to 0.48; the thinner the wire, the closer
the length is to 0.48.
 For thicker wires, a larger segment of the wire has to be removed from
/2 to achieve resonance.

4.7 LINEAR ELEMENTS NEAR OR ON INFINITE PERFECT CONDUCTORS
The presence of obstacles, especially when it is near the radiating element, can
significantly alter the overall radiation properties.
The most common obstacle is the ground. Any energy from the radiating
element directed toward the ground undergoes a reflection. The amount of
reflected energy and its direction are controlled by the ground.
The ground is a lossy medium ( 0) whose effective conductivity increases
with frequency. Therefore it should be expected to act as a good conductor above
a certain frequency, depending primarily upon its composition and moisture
content. To simplify the analysis,
 First assuming the ground is a perfect electric conductor, flat, and infinite.
 The same procedure can also be used to investigate the characteristics of any
radiating element near any other infinite, flat, perfect electric conductor.
The effects that finite dimensions have on the radiation properties of a
radiating element can be accounted for by the use of the Geometrical Theory of
Diffraction and/or the Moment Method.

4.7.1 Imag ge Theor ry
To analyze the performaance of an antenna near an infinite plane conductor,
n a n
vir
rtual sour duced to account for the reflections, which
rces (images) will be introd r
whhen comb bined with the real sources, form an n equivale
ent system
m. The eq quivalent
system give the same radiated field on and a
es above the conduc
ctor as th actual
he
system itself. Below the condu uctor, thee field is zero.

(a) Vertical electric dip
pole (b
b) Field com
mponents at point of ref
flection
Figure 4
4.12 Vertical electric dipole abo
ove an infin
nite, flat, p
perfect elec
ctric condu
uctor

The amount of reflection is generally determined by the respective
constitutive parameters of the media below and above the interface.

For a perfect electric conductor below the interface, the incident
wave is completely reflected and the field below the boundary is zero.

 Vertical polarization
The tangential components of the electric field must vanish on the interface.
Thus for an incident electric field with vertical polarization, the polarization of the
reflected waves must be as indicated in the figure. To excite the polarization of the
reflected waves, the virtual source must also be vertical and with a polarity in the
same direction as that of the actual source (thus a reflection coefficient of 1).
 Horizontal polarization
Another orientation of the source will be to have the radiating element in a
horizontal position, the virtual source (image) is also placed a distance h below the
interface but with a 180 polarity difference relative to the actual source (thus a
reflection coefficient of 1).

In addi
ition to electric so
e ources, artificial equivalent
t“magne
etic”sour
rces and
ma
agnetic co
onductors
s have been introduced.
 Figure 4
4.13(a) displays th source and their imag for a electric plane
he es ges an
conducto The d
or. direction of the arrow id
dentifies the polarity. Sinc many
ce
problems s can be s
solved usiing duality
y.
 Figure 4.
.13(b) illu
ustrates th source and their image when the obstacle is an
he es es t
flat, perfe “magnetic” conducto
infinite, f ect or.

(a) E
Electric con
nductor (b) Magnetic conductor
r
Figure 4
4.13 Electr gnetic sources and th
ric and mag heir images near elec
ctric (PEC) and
magnetic (PMC) cond
m ductors.

4.7.2 Vertiical Electtric Dipoole
Assumi ing a vertical electr
ric dipole is placed a distancce above an infinite, flat,
pe
erfect elec
ctric cond
ductor as sshown in Figure 4.1 12(a).

 For an ob
bservation point P1, there is
s a
rect wave
dir e.

 On the interface, the incid
dent wave e is comp
pletely ref
flected annd the field below
the bounda ary is zero
o. The tanngential c
componen nts of thee electric field mus
st vanish
n the inter
on rface.

1. Radiation pattern

(1) Direct component
The far‐zone direct component of the electric field of the infinitesimal dipole
of length , constant current , and observation point P is given according to
(4‐26a) by

(4‐94)

(2) The reflected component
The reflected component can be accounted for by the introduction of the
virtual source (image), as shown in Figure 4.14(a), and it can be written as

(4‐95, 4‐95a)

(3) The total field
The total field above the interface (z≥0) is equal to the sum of the direct and
reflected components as given by (4‐94) and (4‐95a). In general, we can write that
/ /
2 , 2 (4‐96a, b)

bservation r ≫ h , (4‐96a and (4‐96b) reduce us
For far‐field ob ns a) sing the
bin
nomial exxpansion tto
,    (
(4‐97a,b)

   (4‐98)

2 cos z 0
     (4‐99)
0 0

The shaape and a amplitude e of the field is not t only con
ntrolled b
by the field of the
sin
ngle elem
ment but also by th positio
a he oning of t eleme relativ to the ground.
the ent ve
Th normalized pow patte
he wer erns for 0, /8, /4, 3 /8, /2, and have been
a
plo
otted in F
Figure 4.155..
0 5
15 0 15
0 30 30
-10
0 45 45
-20
0 60 60
-30
0
h=0 5
75 75
-40
0
h=1/8
8 h=3/8
-50
0 h=1/4
4 90
9 h=1/2 90
h=1
-40
0
10
05 105
-30
0
-20
0 120 120

-10
0 135 135
0 150 150
180 16
65 180 165

For h λ/4 more minor lobes, in n addition
n to the m
major one
es, are for
rmed. As
h attains v reater than λ, an even greater nu
values gr n umber of minor lobes is
int
troduced.
.

0 15
0 30
-10 45
These are shown in Figure 4.16 for -20 60
h 2λ and 5λ . In general, the total -30
75
-40
number of lobes is equal to the integer that -50 h=2 90
h=5
is closest to -40
105
-30
2 120
number of lobes 1 -20
-10 135
0 150
180 165

2. Radiation power and directivity
The total radiated power over the upper hemisphere of radius r using
/
1
∙ | |
2
/
| | (4‐101)
which simplifies, with the aid of (4‐99), to

(4‐102)

 As kh → ∞ the radiated power, as given by (4‐102), is equal to that of an
isolated element.
 As kh → 0, it can be shown that the power is twice that of an isolated element.
The radiation intensity can be written as

| | (4‐103)

(4‐103)

The directivity can be written as

4
(4‐104)

The maximum value occurs when kh 2.881 h 0.4585 , and it is equal to
6.566 which is greater than four times that of an isolated element (1.5). The
pattern for h 0.4585 is shown plotted in Figure 4.17 while the directivity, as
given by (4‐104), is displayed in Figure 4.18 for 0 h 5.

Figure 4.17 Elevation plane amplitu
ude pattern
n of a vertica
al infinitesim
mal electric d
dipole at a h
height of
0.4585 ab
bove an infin
nite perfect electric connductor.
Us
sing (4‐10
02), the radiation re
esistance can be written as

| |
2 (4‐105)

(4‐19)

Th radiation resista
he 4.18 for 0 h
ance is plotted in Figure 4
p 0 5 when = /50
5
an
nd the element is ra
adiating innto free‐s
space (η  120).

Figure 4.18 Directivity and radiation
i D n n resistance of a vertical infinitesimal electric d
dipole as a fu
unction of
its height above an infinite perfectt electric conductor

3. monopo
ole
In prac
ctice, a w
wide use has been made o a quar
n of rter‐wavelength m monopole
( λ/4) m
mounted above a g
a ground plane, and fed by a coaxial line, as s
d a shown in
Fig
gure 4.199(a). For analysis purposes a λ/4 image is introduc
s, ced and it forms
the λ/2 eq quivalent of Figur 4.19(b). It should be e
re emphasize that the λ/2
ed
eqquivalent of Figure 4.19(b) g gives the correct field value
es for the
e actual sy ystem of
gure 4.19(a) only above the interface
Fig e (z 0, 0 θ /2).

gure 4.19 Quarter‐
Fig ‐waveleng
gth monopole on a
an infinite perfect e
electric co
onductor

Thus, the far‐zone electric and magnetic fields for the λ/4 monopole above
the ground plane are given, respectively, by (4‐84) and (4‐85).

, (4‐84, 4‐85)

The input impedance of a λ/4 monopole above a ground plane is equal to
one‐half that of an isolated λ/2 dipole. Thus, referred to the current maximum,
the input impedance Z is given by
Z monopole Z dipole 73 j42.5 36.5 j21.25 (4‐106)

4.7.4 Antennas for Mobile Communication Systems
 The dipole and monopole are two of the most widely used antennas for
wireless mobile communication systems.
 An array of dipole elements is extensively used as an antenna at the base
station of a land mobile system while the monopole, because of its broadband
characteristics and simple construction, is perhaps to most common antenna
element for portable equipment, such as cellular telephones, cordless
telephones, automobiles, trains, etc.
 An alternative to the monopole for the handheld unit is the loop. Other
elements include the inverted F, planar inverted F antenna (PIFA), microstrip
(patch), spiral, and others.
The variations of the input impedance, real and imaginary parts, of a vertical
monopole antenna mounted on an experimental unit are shown in Figure 4.21.

Figure 4.21 Input impedance, real and i
imaginary parts, of a ve
ertical mono
opole mount
ted on an
expe
erimental ce hone device
ellular teleph e.

It is a
apparent that the first reso
onance, around 1,0 MHz, is slowly varying
000 y
values of immpedance versus frequency and of desirable magnitude, for practical
e y, f

im
mplementa
ation.
Above the first
t resonance, the im
mpedance
e is induct
tive. The s
second re
esonance
rapid changes in th values of the impedance. These values and variation of
he s e
im
mpedance are usually undesirable for practical implementation.

4.7
7.5 Horizo
ontal Elec
ctric Dipo
ole

When the line eleme
n ear ent
is placed ho
orizontally
y relative
e to
the infinitte electric grou und
plaane, as sh
hown in Fiigure 4.24
4.

Fig
gure 4.24 Ho
orizontal eleectric dipole,, and its associated
imaage, above a
an infinite, f
flat, perfect nductor
t electric con

The aanalysis p
procedure of this is identica to the one of th vertica dipole.
e s al he al
Int
troducing an imag and assuming far field observat
g ge a tions, as shown in Figure
4.2
25(a, b),

(a) Horizontal electric
c dipole abo
ove ground p
plane (b) Far‐
‐field observ
vations
gure 4.25 Ho
Fig orizontal ele
ectric dipole
e above an infinite perfe
ect electric conductor
oefficient is equal to R
Since the reflection co 1, The direct and the
ref
flect components c can be wr ritten as
(4‐111)

⟹ (4‐112)
ind the angle ψ , which is measu
To fi ured from the y‐
m ‐axis tow
ward the
ob
bservation
n point, w
we first for
rm

∙ ∙ (4‐113)
⟹ 1 1 (4‐114)
Since for far‐field observations

for phase variations (4‐115a)

for amplitude variations (4‐115b)
the total field, which is valid only above the ground plane (z≥h; 0≤θ≤/2, 0≤
≤2), can be written as

E 1 sin sin 2 sin cos (4‐116)
Equation (4‐116) again consists of the product of the field of a single isolated
element placed symmetrically at the origin and a factor (within the brackets)
known as the array factor.

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