Bx26501512

A. Alwadie / International Journal of Engineering Research and Applications
(IJERA) ISSN: 2248-9622 www.ijera.com
Vol. 2, Issue 6, November- December 2012, pp.501-512
High Performance Predictive Direct Torque Control Of Induction
Motor Drives System
A. Alwadie
Electrical Engineering Department, Faculty of Engineering, Najran University, KSA

ABSTRACT
This paper presents a practical and DSP board) are implemented and configured.
implementation for direct torque control of In addition, Implementation of Digital Signal
induction motor drive. Control system Processor (DSP1102) on DTC has been presented.
experiment is proposed using Digital Signal Through the implementation, it will be shown that
Processor. This control scheme directly the high performance of the control system is
determines the switching states of the inverter achieved. Computer simulation validates the
and gives optimal characteristics for stator flux experimental results of this control method.
and torque control. A simple switching pattern
for direct torque control algorithm for induction 2. DIRECT TORQUE CONTROL MODELING
motor drives has been presented. The main A detailed block diagram of the direct
results with the simple approach are greatly torque controlled induction motor drive is shown in
reduced torque pulsations, constant and Fig.(1). The figure shows a quick response of the
controlled switching frequency. Results show that torque control system in which the instantaneous
the simulation meets the experimental results. speed (s) of the primary flux linkage vector (ys)of
the motor is controlled in such a way that it allows
Keywords - Digital Signal Processor (DSP) - the developed torque (Td) to agree with the torque
Induction Motor (IM) - Direct Torque Control command (T*) within a certain error. ys and Td are
(DTC) - Space Vector Modulation. calculated respectively by the following relations
[4]:
1. INTRODUCTION
Nowadays, field oriented control (FOC)    (v s  R s i s ) dt (1)
s
and direct torque control (DTC) is amongst the most
p i  i 
3
T   ds qs qs ds 
important control methods and is extensively used in (2)
induction motor drives. High performance induction
d 2  
motor drives without position sensors are desired for
industrial applications. From this point, DTC is a Where,
suitable technique for squirrel-cage induction Vs the instantaneous space primary voltage vector.
motors. In this technique, the stator flux is is the instantaneous space primary current
implemented by measuring the stator voltage and vector.
currents in a similar way that is done in direct FOC Rs the primary winding resistance.
technique. On the other hand, the correct choice of yds, yqs d-axis and q-axis stator flux.
the inverter voltage vector leads to direct and ids, iqs d-axis and q-axis stator current.
decoupled control of motor flux and torque.
According to this novel concept of decoupling DTC From equation (1), it is clear that ys can be
is seen to be different from FOC and it has satisfied directly controlled by vs because of small stator
some advantages like simplicity and quick torque winding resistance drop. It is generally known that
response [1:3]. there are eight voltages made by a three-phase
In this paper, a control method is proposed inverter. By proper selecting of these vectors both
that can provide transient performance similar to the amplitude and speed of ys are controlled to
DTC with reduced torque pulsations at steady state. regulate the instantaneous value of Td as it will be
The inverter switching pattern is determined over shown later.
constant switching period and hence constant
switching frequency is obtained. The torque and flux 2.1 Control system configuration
controller design is investigated using a sliding For the purpose of control design, it is
mode controller for direct torque control. Moreover, necessary to know a dynamic model of the induction
the complete power circuits such as (Power supply, motor, which incorporates all dynamic effects
Rectifier, pulse width modulation (PWM) inverter, occurring during steady state and transient operation.
power interface, current sensors, voltage sensors, The state equations of the induction motor are given
by [5]:

501 | P a g e


* Speed
 - + controller

Injected signal
Torque Comparator
*
+ T
Id, iq Calculation Td - - +
Voltage vector
of torque selection tables
Calculation of s 6 1 2
,
sd sq 3
5 4
Calculation of  -
s
stator flux
Vd, vq * +
s
Sa
Flux
d-q transformation of Ia, Ib Comparator Sb
currents and Sc
voltages Va, Vb

IM Ed 3phase-supply

Incremental

encoder Inverter Rectifier

Fig.(1) Schematic diagram of direct torque control system

 dids   1 Vds 
 dt   Ls 
 di    Lm mutual inductance per phase referred to the
1
 qs   Vqs  stator.
 dt    Ls  S total equivalent leakage factor of the motor.
 didr   Lm +
 dt   Vds 
Ls Lr  L2 
 di      1  m 

 qr
 

Lm   Ls Lr 

 dt   L L qs 
V
   s r  Ls stator winding inductance per phase(Ls
=Lls+Lm).
 Rs Lm p r
2
Rs Lm Lm p r 
  Lr rotor winding inductance per phase(Lr=
 2 Ls  Ls Lr Ls Lr Lr   Llr+Lm).
 Lm p r Rs L p r Lm Rs  ids 
 L L   m  
Ls Lr Ls Lr  iqs  The instantaneous space vectors of the
 s r
  PWM inverter are regarded as discrete values, and
 Rs Lm Lm p r R  i
  r  r   dr  the analysis using the instantaneous vectors is
 Ls Lr Lr Lr   iqr 
 L p   suitable for investigating the dynamic behavior of
Lm Rs r Rr  the motor. To treat three phase quantities as a whole,
 m r
 
 Lr
 Ls Lr  Lr  
the three phase machine voltages and currents are
represented by an instantaneous space voltage vector
(3)
(vs) and an instantaneous space current vector (i s),
where,
respectively, as [6]:
Lls stator winding leakage inductance per
phase. 2 
 j 2 j 4  (4)
v 
s v v e 3 v e 3 
a c
Llr rotor winding leakage inductance per phase. 3  b 
 

502 | P a g e


2  j 2 j 4  (5)
is  ia  i e 3  ic e 3 
3  b 
 

where, va, vb and vc are the instantaneous values of
the primary line-to-neutral voltages and ia, ib and ic
are the instantaneous values of the primary line-to-
neutral currents.

a Coupling

Ed sa s sc
n DC
b
Stat b c Gen.
or

Fig.(2 ) Schematic diagram of PWM inverter fed induction motor drive.

Schematic diagram of the PWM fed induction motor d-axis
V1(1,1,0) V2 (0,1,0)
drive is shown in Fig.(2), where Sa, Sb, and Sc are the
switching functions with the value of 1 when the
switch is set to positive voltage or 0 when the switch
is set to negative voltage.
Considering the combinations of the status
V6(1,0,0) V7(1,1,1) V8 (0,0,0) V3(0,1,1)
of switches, the inverter has eight conduction q-axis
modes[7,8]. By using these switching functions the
primary space voltage vector can be expressed as:
 2 4
j 
 
v s Sa , Sb , Sc 
2
3
Ed  Sa  Sb e 3  Sc e 3  (6)

j
 V5(1,0,1) V4(0,0,1)
where, Ed is the dc link voltage of the Fig.(3) Representation of primary space voltage
inverter. vectors.
According to the combinations of switching modes,
the primary space voltage vectors vs(Sa, Sb, Sc) is 2.2 DTC STATE VARIABLE SELECTION
specified for eight kinds of vectors vs(1,1,1), (0,0,0) The direct torque control is achieved using
and the others are the space nonzero active voltage three state variables [9]. The first variable is the
vectors. i.e.(1,0,0),………..(0,1,1), as shown in difference between the command stator flux and the
Fig.(3) and according to table(1). estimated stator flux magnitude, the second state
Table (1) variable is the difference between the command
N(switching mode) 1 2 3 4 5 6 7 8 torque and the estimated electromagnetic torque, and
phase-a 1 0 0 0 1 1 1 0 the third one is the angle of stator flux. The control
phase-b 1 1 1 0 0 0 1 0 variables of the direct torque control can be
investigated using Fig.(4-a), Fig.(4-b) as follows.
phase-c 0 0 1 1 1 0 1 0
Looking at the flux position in Fig (4-a) it is noticed
that, states 3,4,5 will increase the flux while states
2,1,6 will decrease it. Similarly, states 4, 3, 2 will
increase the current (i.e. the torque) while states
1,6,5 will decrease it. Gathering these states, it is
found that for a large increase in flux and a small
increase in torque, state 4 is selected, for a small
increase in flux and a large increase in torque, state 3
is selected, for a small decrease in flux and a small
increase in torque, state 2 is selected, for a large
decrease in flux and a small decrease in torque, state
1 is selected, for a small decrease in flux and a large
decrease in torque, state 6 is selected, for a small
increase in flux and a large decrease in torque, state

503 | P a g e

5 is selected, and for a small decrease in torque and a (2N-3)p /6 = <fn<= (2N-1)p /6 where;
constant flux, state 0 is selected. N=1,2,……6.

3. SLIDING MODE DIRECT FLUX AND TORQUE Since the flux is kept almost constant, the
CONTROLLER DESIGN torque is then proportional to s (instantaneous-slip
In order to achieve a high level of ac motor angular velocity). As mentioned above, the velocity
control, it is necessary to keep the flux as nearly as of the primary flux is proportional to the dc link
constant except for the field weakening region [3,4]. voltage, and the direction of rotation of the primary
In case of sinusoidal ac power supplies the trajectory flux is determined by the active voltage vectors and
of primary flux is a smooth circle unlike, it happens the movement of the primary flux will be stopped
in electronic inverters, which is not purely when zero voltage vectors is applied. Therefore, by
sinusoidal. Therefore, it is difficult to keep the air changing the ratio of the number of active and zero
gap flux constant especially in the low speed region. voltage vectors, the slip angular velocity can be
The instantaneous space vector approach has made it regulated. Hence, the direct torque control is
possible to trace the primary flux and keep it almost obtained by adequate selection between active and
constant. The primary space flux vector of induction zero voltage vectors. Hence, when the torque Td is
motor can be calculated by equation (1). Assuming small compared with T* it is necessary to increase Td
that the voltage drop of the primary winding as fast as possible by applying the fastest s. On the
resistance is small, the trajectory of ys moves in the other hand, when Td reaches T* it is better to
direction of vs (Sa, Sb, Sc), the velocity of space non decrease Td as slowly as possible. Thus the output of
zero voltage vector gives the direction and the the torque controller can be classified as follow:
amplitude of ys , but by applying a space zero T=1 if Td < T*
voltage vector to an induction motor, the movement T=0 if Td= T*
of ys will be stopped. Therefore, by selecting these (8)
Vs dI2

Is dIdI1
6 dI3
dI4
Is dI5

q q

s
d d
d d d
d d
d
(a) (b)

Fig.(4) Vector diagram illustrating the state selection process of flux and torque during transient operation.

vectors appropriately, the trajectory of ys will follow T=-1 if Td>T*
the specified locus. i.e. by selecting adequate voltage The output of the torque controller is a three level
vectors, ys can be kept almost constant and the hysteresis comparator. Using the three variables, y,
rotating velocity of ys can be controlled by changing T and the stator flux position f =(tan -1 y d / y q), the
the output ratio between zero vectors and the following optimum switching tables (2), and (3) are
remaining non zero vectors. So, s and ys are greatly obtained. Accordingly, accessing the tables, in
concerned with the direct torque control. In DTC which the inverter output voltages are given by the
system under study, constant flux can be obtained by output of the two comparators and the angle f(n), the
comparing the primary flux with the command flux. optimum switching model (base drive signals)
The output of the flux comparator will be as follows: desirable for driving the base of the inverter can be
y = 1 when y s < y s*., and obtained.
y = 0 when y s > y s*. The inverter output voltage is then defined as in
where, e denote the error signal in both y s and Td. equation (6). The entire system has been verified
using Matlab, Simulink and the following simulation
In the six step inverter, the d-q plane can be divided results has been obtained.
into six regions resulting from the general case:

504 | P a g e

Table (2)  Stator flux estimation using motor speed and
 T 1 2 3 4 5 6 two line currents (current model)
0 1 V2 V3 V4 V5 V6 V1 One way to eliminate the drift is to use the rotor
model in stator coordinate equation (9) in which the
0 0 V8 V7 V8 V7 V8 V7
required signals to be measured are stator currents
0 -1 V4 V5 V6 V1 V2 V3 and rotor speed.

Table (3) d r Rr * Lm * is R
 T 1 2 3 4 5 6   ( r  j ) (9)
dt Lr r L r
1 1 V1 V2 V3 V4 V5 V6 r
1 0 V7 V8 V7 V8 V7 V8 Lm  2 L
  i (L  )  m  (10)
1 -1 V5 V6 V1 V2 V3 V4 s s s Lr Lr r

3.1 STATOR FLUX ESTIMATION ANALYSIS The accuracy of this estimator depends on
In order to have a fast acting and accurate the correct setting of the motor parameters
control of induction motor, the flux linkage of the particularly on the rotor time constant which is a
motor must be measured. However, it is difficult and critical and the mutual inductance which is less
expensive. Therefore, it is preferred to estimate the critical. This estimator has good properties at low
motor flux linkage based on measurement of motor speeds; however, it is sensitive to parameter errors at
voltages, currents and speed if it is required. There high frequency.
are several flux estimation techniques, some of them
work with stator flux while others prefer rotor flux.  Stator flux estimation using flux observer
As stator flux control is used in the experiment, the As the voltage model estimator has good
estimators discussed here are mainly stator flux properties at high frequency and the current model
estimators. Two cases are considered throughout this estimator has good properties at low frequency,
study: first, neglecting the rotor current while the various ways have been tried to combine them. A
motor is running at no load, the second is to consider straightforward way of combination is adopted [4]
the rotor current during load condition. using a low pass filter where the current model
estimator and the voltage model estimator are
 Stator flux estimation using dc link voltage selected at low and high frequency, respectively.
and two line currents (voltage model) The same idea of combining the voltage and current
One desirable feature of direct torque models to obtain a flux estimator at all speeds is
control, or any control method based on stator flux observed in [8]; however, instead of using a low pass
control is to consider usually the stator flux as an filter to combine the models an observer topology is
estimated quantity using terminal measurements [6]. created which uses a closed loop linear controller
At first, simple flux estimator is obtained from with two deterministically set gains to govern the
equation (1) in integral form which is known as transition between the low and high speed
voltage model [7]. The input signals to this estimator estimators. However, the drawback of this type of
are measured stator currents and voltages and if observer is that they require a speed or position
these signals are without errors like offset, noise and information using a shaft sensor and if the speed
if the stator resistance is well tuned, then this information can be obtained without sensor then a
estimator will be a perfect solution even if the rotor wide speed range flux estimation could be achieved.
current is considered not to be zero. In fact this
estimator is difficult to be used in practice, because The attractive feature of stator flux
of the acquired error signals, offset and drift in the regulated, rotor flux oriented control technique is
integrating hardware. As a result, accumulation of that it is capable of achieving correct torque and flux
these errors will exist if there is no feedback from control even under detuned conditions [9]. It is clear
the integrator output to the input. The resulting run that the performance of flux estimation depends on
away is a fundamental problem of pure integration. voltages, currents and the model parameter
However, the challenges in stator flux estimation matching.
appear at low frequencies especially when the stator
resistance drop is in the same order of the stator 3.2 IMPROVED STATOR FLUX ESTIMATION
voltage. The problems related to drifting integrators Induction motor stator flux estimation has
are often pointed out as the main difficulties with taken a lot of interest as it is used with several
flux estimation at low speeds and a solution to stator control techniques, such as, classical DTC, stator
flux estimation will be presented later. flux field oriented control. Even though Stator flux
can be measured [10], this is really difficult and
expensive; consequently, the flux is preferred to be
estimated based on measurement of voltages,

505 | P a g e

currents and velocity if it is required. Several synchronous frame flux regulator requires d-q to a-
problems are still under investigation like zero speed b-c additional transformation and the angle of
operation without speed sensor. The main reasons transformation can be calculated as follows, As a
related to the voltage model are offset and drift simple solution to delete d-q to a-b-c additional
component in the acquired signals and the increased transformation and then the control algorithm will be
sensitivity to stator resistance variation. These nearly similar to standard DTC which has the
reasons decrease the accuracy of stator flux advantage of no reference frame transformation. The
estimation around zero speed. Other difficulties motor slip angle will be calculated through torque
arises if the dc link voltage and the switching signals comparator and PI controller as shown in Fig. (1).
of the inverter are used to calculate the stator flux, The angle of the stator flux can be obtained easily in
one is the voltage drop across the power devices stator reference frame using the rotor position
themselves, the other is the voltage losses due to through a position sensor or a sensor less solution in
dead time between two inverter switches in the same addition to the slip angle which has been calculated.
branch. The exact voltage calculation is difficult
with respect to the dead time, but these two errors 4. SIMULATION RESULTS
must be compensated. A simple solution is to use The developed control system has been
low pass filter to avoid integration drift [12], implemented in a computer simulation program,
however, this results in magnitude and phase error of modeling the two subsystem induction motor and
the estimated flux. Implementing a pure integrator control algorithm. The controller simulation uses the
avoids the bandwidth limitation associated with low parameters of an experimental laboratory prototype.
pass filter, especially at low frequencies [13]. These parameters are listed in the Appendix. The
The modified block diagram of stator flux vector graphs shown in Fig.(5) depict the response of the
control scheme is represented in Fig. (1) and the system with a torque command of 4 N.m. The
motor torque is expressed as a cross product of the transient and steady state flux vector in Fig.(5-a)
stator and rotor flux as in equation(11). shows nearly a circular path indicating a good flux
3 Lm   regulation. The electromagnetic torque shown in
T  P
d 2 Ls Lr  r . s   r . s  
  Fig.(5-b) is close to the commanded value, while the
 
time taken for the torque reach its commanded value
(11) is about 0.02 Sec., ensuring good dynamic torque
and the motion equation is as follow: response. Fig.(5-c) shows the stator currents in phase
T  TL dr a, b and c which are nearly sinusoidal. Fig.(5-d)
e  P  B * r (12)
shows the waveforms of stator currents in d-q axis, it
J dt
is noticed that the phase shift between them equal
Where: P is the number of pair poles, J is the 900.
combined rotor and load inertia coefficient,  is the Figure(6) shows the response of the system
combined rotor and load friction coefficient. In for a step change in torque from 3 N.m. to 7 N.m.
similar to rotor field orientation the stator flux keeping the flux command constant. Fig.(6-a) shows
components can be written in synchronous reference the steady state stator flux, which demonstrates an
frame as follow: excellent dynamic flux control under changed
Ls command torque. Fig.(6-b) shows the developed
 *e  * (1   Tr * P) λ* (13) torque which is enclosed to a commanded value of 3
ds Lm r
N.m. region or 7 N.m. region. The three phase stator
currents are shown in Fig.(6-c). It is clear that, these
2LsLr Te *
 *e qs  * (14) stator currents are changed at the instant of changing
3PLm r * the command torque and swiftly retrieve their steady
state values.
In equation (4) it is clear that a change in
the control variables yds in synchronous reference
frame is followed with a time constant Tr which is
much lower than the rotor time constant Tr involved
in traditional stator current control and this gives the
facility to faster controller. From equation (5) it is
noticed that with constant rotor flux command a
change in yqs is followed by a corresponding change
in torque and this will give decoupling. In principle,
the control method proposed here is based on driving
the stator flux vector toward its reference vector by
applying the appropriate voltage vector and a normal
PWM method to obtain constant switching
frequency and reduced torque pulsations. The

506 | P a g e


qs qs

ds
(a) Stator flux. ds (a) stator flux.

(b) Electromagnetic torque. (b) Electromagnetic torque.

(c) Stator currents.
(c) Stator currents.
Fig. (6) System response for step change in load
torque.

Figure(7) shows simulation results in which a step
response of the system is studied at a step change of
command torque from 3 N.m. to its 7 N.m. at 0.4
Sec., then, at 0.8 Sec. the command torque is
decreased to its initial value (3 N.m.). Behavior of
developed torque shows clearly that the response of
the torque is as quickly as possible as shown in
Fig.(7-b). As can be seen from Fig.(7-c), it is clear
that, the stator currents in phases a, b, and c are
changing according to the variation of the command
torque. The values of currents in the region, where
(d) d-q axis stator current. the torque is increased to 7 N.m, is increased to 3.5
Fig. (5) System response at load torque 4 N.m. and Amp.(r.m.s.).
motor speed 1500 rpm.

507 | P a g e

5. EXPERIMENTAL IMPLEMENTATION
Several dynamic tests were performed to
investigate the validity of the proposed control
method. The experimental set-up used for the
practical investigation is shown in Fig.(8). The
induction motor under test is coupled to a self-
excited DC generator. The motor is energized from
an inverter bridge, which consists of six IGBTs in
conjunction with stepper circuits. The DC-link is
provided from three-phase bridge rectifier. An
incremental encoder is coupled to the motor shaft to
measure the motor speed. Actual stator currents are
measured using current sensors. The stator voltages
are sensed using voltage transducers. This system is (b) Electromagnetic torque.
fully controlled by a DSP controller board which is
installed on a PC computer. Fig. (7) System response for step change in load
torque.
A series of experimental results was carried
out to verify the validity of the proposed control
scheme for the drive system. The results were
obtained at different operating points. The results
were obtained at different operating points. First, the
drive system response was obtained at heavy load
and a speed below the base speed. Figure (9) shows
qs the motor speed response at 1200 rpm. The motor
started with almost no overshoot and follows its
command with nearly zero steady state error. The
motor phase currents are also shown in Fig. (9).
The response of the control system due to a
step change in speed command is shown in Fig. (10).
The speed command is changed from 120 rad/s to
ds 100 rad/s. It can be seen from Fig. (10) that the
(a) stator flux. motor speed is decelerated smoothly to follow its
reference value. Figure (11), show the motor
response for step up in speed command, which
change from 100 rad/s to 150 rad/s.

508 | P a g e

3-phase
AC supply Edc Sa Sb Sc

Load
Sa Sb Sc

Incremental
encoder

Gating drives
IM DC Gen.

Oscilloscope
Vector Control Gate Pulse
Generator
D/A A/D

TMS320C31
Converte Converte
r r
Incremental
Data-bus encoder
RS-232C
interface

Host Computer DSP board (ds1102)

Fig. (8) Experimental set-up for DSP-based control of IM drive

Fig. (9) Drive response for reference speed of 1200 rpm (x-axis Time (ms))

509 | P a g e


Fig. (10) System performance for step down in speed command (x-axis Time (ms))

Fig. (11) System performance for step up in speed command (x-axis Time (ms))

510 | P a g e

6. CONCLUSIONS Components and Systems Journal, Vol.
In this paper, the basic concept of DTC is 34, No. 3, pp. 303-319, March 2006.
implemented using a DSP control system by [7] A. Linder: “ A Rapid-Prototyping System
modifying the classical method. The advantage of Based on Standard PCs with RTAI as
the proposed method lies in achieving constant Real-Time Operating System, ” Third
switching frequency and solving the starting Real-Time Linux Workshop, Milan, Italy,
problem without a dither signal. A further November 26 -29, 2001
improvement of this work can be achieved by using [8] R. Kennel, A. El-refaei, F. Elkady, E.
stator flux estimation and reduced torque pulsation Elkholy and S. A. Mahmoud “Improved
techniques. The existence of non linear torque and direct torque control for induction motor
flux hysteresis controllers result in noticeable drives with rapid prototyping system”
torque pulsations and variable switching frequency. IEEE, IECON, 2003 U.S.A.
Thus, the proposed control method can give [9] R. Kennel, A. El-refaei, F. Elkady, E.
transient performance like DTC and reduced torque Elkholy and S. A. Mahmoud “ Torque
pulsations at steady state. An inverter switching Ripple Minimization for Induction Motor
pattern that can achieve DTC method for induction Drives with Direct Torque Control (DTC),
motors in which the analytical derivation is based ”The Fifth International Conference on
on the prediction of the voltage vector direction Power Electronics and Drive Systems
which is required to compensate electromagnetic PEDS 2003, Singapore, November 17 –
torque and stator flux errors. In general, an 20, 2003.
optimistic experimental results and an improved [10] J. Chen. Yongdong “ Virtual Vectors
performance over other existing schemes is Based Predictive Control of Torque and
obtained. Flux of Induction Motor and Speed
Sensorless Drives, ” IEEE, IECON 1999.
REFERENCES [11] Y. Xue, X. Xu, T. G. Habetler and D. M.
[1] J. Holtz and J. Quan “ Sensorless Vector Divan “ A Stator Flux-Oriented Voltage
Control of Induction Motor at Very Low Source Variable-Speed Drive Based on dc
Speed Using a Non Linear Inverter Model Link Measurement,” IEEE Transaction
and Parameter Identification,” IEEE Industry applications, Vol. 27, No. 5, pp.
Transaction Industry Applications Society 962-969, September/October 1991.
Annual Meeting, Chicago, Sept. 30- Oct. [12] T. G. Habetler, F. Profumo, G. Griva “
4, 2001. Stator Resistance Tuning In A stator Flux
[2] M. Depenbrock, “ Direct Self Control Field Oriented Drive Using An
(DSC) of Inverter-fed Induction Motors, ” Instantaneous Hybrid Flux Estimator, ”
IEEE Transaction Power Electronics, Vol. The European Power Electronics
3, No. 4, pp. 420-429, October1988. Association, Brighton, No. 13, pp. 292-
[3] I. Takahashi and T. Noguchi “ A new 299, September, 1993.
Quick Response and High Efficiency [13] N. R. Idris, A. Halim, N. A. Azly, “ Direct
Control Strategy of An Induction Motor, ” Torque Control of Induction Machines
IEEE Transaction Industry Applications, With Constant Switching Frequency and
Vol. IA-22, No.5, PP. 820-827, .Sept./Oct. Improved Stator Flux Estimation,” 27th
1986. Annual Conference of the IEEE Ind. Elect.
[4] E. Flach, R. Hoffmann, P. Mutschler “ Society, pp. 1285-1291.
Direct Mean Torque Control of Induction [14] L. Jansen, D. Lorenz “Observer Based
Motor, ” in Conf. Rec. of the European Direct Field Orientation Analysis and
Conference on Power Electronics and Comparison of Alternative Methods,”
Applications, pp. 3672-3676, 1997. IEEE Transaction Industry applications,
[5] E. E. El-Kholy , S. A. Mahmoud, R. Vol. 30, No. 4, pp. 945-953, July /August
Kennel, A. El-Refaei, and F. El-Kady, 1994.
“Torque Ripple Minimization for [15] B. Kenny, D. Lorenz “ Deadbeat Direct
Induction Motor Drives With Direct Torque Control Of Induction Machines
Torque Control” Electrical Power Using Self Sensing at low and zero speed,
Components and Systems Journal, Vol. ” Ph. D thesis Wosconsin, Madison, U. S.
33, No. 8, pp. 845-859, August 2005. A., 2001.
[6] E. E. El-Kholy , S. A. Mahmoud, R. [16] J. Holtz, J. Quan “ Sensorless Vector
Kennel, A. El-Refaei, and F. El-Kady, Control of Induction Motors at Very Low
“Analysis and Implementation of a New Speed, ” Ph. D thesis, Wuppertal
Space-Vector Current Regulation for university, 2002.
Induction Motor Drives” Electrical Power [17] A. B. Plunkett, J. D. D’atre, and T. A.
Lipo,“ Synchronous Control of Static AC

511 | P a g e

Induction Motor Drive, ” IEEE Trans. On
Industry Applications, Vol. IA-15, No.4,
pp. 430-437, July/August 1979.
[18] D. Casadei, G. Serra, A. Tani, L. Zarri “
Performance Analysis of a Speed
Sensorless Induction Motor drive Based
on a Constant Switching Frequency DTC
Scheme, ” IEEE, IECON, 2000.
[19] Habetler “Direct Torque control of an
Induction Machines Using Space Vector
Modulation,” IEEE Transaction Industry
Applications, Vol. 28, No. 5, pp. 11045-
1053, September/October 1992.
[20] J. Monteiro, G. Marques, and J. Palma “A
Simplified Stator Flux Vector Control
Method in d-q Co-ordinates For Induction
Machines,” ISIE 2000.

I. Appendix
The induction motor is a squirrel cage and has the
following data:
Rated power : 1.1 KW
Rated line voltage : 380 volt.
No. of pole pair : 2.0
Stator resistance : 4. 4826 ohm.
Rotor resistance : 3. 6840 ohm.
Stator leakage inductance : 0.0221 Henry.
Rotor leakage inductance : 0.0221 Henry.
Mutual inductance : 0.4114 Henry.
Supply frequency : 50.0
Rated load torque : 11.0 N. m.

512 | P a g e

Bx26501512

Recommended

Recommended

More Related Content

What's hot

What's hot (20)

Viewers also liked

Viewers also liked (9)

Similar to Bx26501512

Similar to Bx26501512 (20)

Bx26501512